Digital radar or sonar apparatus

ABSTRACT

A radar or sonar system amplifies the signal received by an antenna of the radar system or a transducer of the sonar system is amplified and then subject to linear demodulation by a linear receiver. There may be an anti-aliasing filter and an analog-to-digital converter between the amplifier and the linear receiver. The system may also have a digital signal processor with a network stack running in the processor. That processor may also have a network interface media access controller, where the system operates at different ranges, the modulator may produce pulses of two pulse patterns differing in pulse duration and inter-pulse spacing, those pulse patterns are introduced and used to form two radar images with the two images being derived from data acquired in a duration not more than twenty times larger than the larger inter-pulse spacing, or for a radar system, larger than one half of the antenna resolution time. One or more look-up tables may be used to control the amplifier. The radar system may generate digital output which comprises greater than eight levels of radar video.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.12/673,796 filed Jan. 10, 2011, which is a 371 national stageapplication of International Patent Application No. PCT/GB2008/002935filed Aug. 29, 2008, which claims priority to GB Patent Application No.0717031.9 filed Aug. 31, 2007, which are all incorporated herein byreference in their entirety as part of the present disclosure.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a radar or sonar apparatus. It isparticularly concerned with such an apparatus including digitalprocessing of the radar or sonar signals received

2. Summary of the Prior Art

Radar scanners have been manufactured since the 1940's. The developmentof radar has diverged into two methods, non-coherent pulse radars andDoppler radars. Doppler radars have received considerable developmentfor military and aviation use. Modern Doppler radars make extensive useof digital signal processing and digital control. However pulse radarshave remained extensively analog, using logarithmic receivers and analoggain controls to generate baseband video that is sent to the display forprocessing. The display converts the baseband video to digital formusing a 1-bit comparator. Any signal processing that takes place in suchdisplays is limited by the thresholding that take place in thecomparator. Analogue processing and control has hereto been cheaper andadequate for most purposes.

Radar receivers have historically been based upon down conversion fromS-band or X-band to an intermediate frequency (IF) of the order of 60MHz. The signal at the intermediate frequency (IF) is then subject to IFfiltering and analog logarithmic demodulation to baseband “video”,followed by baseband filtering. The radar signal processor is oftenplaced in radar display apparatus, rather than the scanner itself. Thisis because there are size, weight and power constraints on the radarscanner enclosure. The scanner is exposed to the worst of theenvironment, whereas the display enclosure is often placed in a morebenign environment such as a heated cockpit. When multiple displays areused, the radar information is typically sent out over a network, theradar signal processors, in the each display apparatus not connecteddirectly to the scanner, are idle and redundant.

Analog IF filters are used for conventional mono-pulse radars. Theoptimum bandwidth for the IF filter (BW)=l/T, where T is thepulse-width.

The pulse-width is optimised for range resolution and average power.Short pulses are used for short range, where the highest rangeresolution is needed and the return signals are strongest. Long pulsesare used when the absolute range resolution can be reduced, but highertransmit power is needed due to the lower power of the return signal.

The returned signal power is:s=k/R^4Where k is a constant and R is range

The range resolution of the display limits the displayable rangeresolution to a maximum proportion of the range say 1/1024. Thus atlonger ranges the absolute range resolution can be reduced. It should benoted that increasing the range resolution reduces clutter; the excessnumber of range bins can be combined in the signal processor prior todisplay. Pulses of this optimum transmitted pulse-width are selectedwhen the user changes instrumented range.

When a large number of different pulse-widths (say 8) are available,providing matched analog filtering becomes onerous to set-up, and liableto drift. Hence it is normal to provide a compromise sub-set of filterbandwidths, and use each one to cover several pulse widths. This leadsto sub-optimal filtering. Analog video filtering follows detection toremove the spurious products that arise from the use of a logarithmicdetector. The analog video loss-pass filter works at baseband and isrequired to have wide bandwidth to cut-off frequency ratio, yet havelinear phase, so video filtering is minimal.

The radar signal requires conversion from analog to digital form usingan Analog to Digital Converter (ADC) or more often just a comparator.The signal is required to be in digital form for use with a rastergraphic display such as a CRT or LCD. This is because it has to be(scan) converted from polar (Range, Azimuth) to Cartesian co-ordinates(x, y). Prior to conversion low-pass video filtering is required toavoid aliasing artefacts. The log processing of the amplitude from aconventional radar receiver is a non-linear process, the output of whichcomprises many harmonics that have to be removed to avoid aliasing inthe ADC. However as discussed above, the video filtering is minimal. Thepoor removal of these spurious products causes aliasing, and destroysinformation that could have been used in later signal processing, forenhancing the quality of the displayed information. The logarithmicbaseband video precludes any signal processing that makes use of linearsignals for example, Fast Fourier Transforms (FFTs).

Logarithmic detector receivers have historically had advantages in radarreceivers due to their large dynamic range and are inherently CFAR(Constant False Alarm Rate). To ensure that these qualities are notcompromised in a linear receiver, the dynamic range of the analog andanalog-to-digital conversion must be very wide. Low noise pre-amplifiercomponents, a high speed, high resolution (large numbers of bits)analog-to-digital conversion, very fast digital filters and ideally afloating-point processing digital signal processor (DSP) must be used.These have been expensive and difficult to use. Thus linear detectionand processing have net previously been considered, being bothuneconomic and unreliable (due to high power dissipation).

Dual range scanners are available, which allow the display of separateradar plan position indicators (PPIs), which are the conventional radardisplays. However, in such scanners the scanner makes more than one fullrotation using each pulse type, with the receiver optimised for thereception of one pulse type at a time. The displays from each range havethe disadvantage of clearly not being simultaneously updated, whichcreates an ambiguity. Time is allowed for the receiver and transmittercharacteristics to be changed, resulting is a period when no pulses aretransmitted or received on any range. There is a resulting loss ofinformation.

FIG. 1 of the accompanying drawings shows a known radar apparatus. Itcomprises five principal components, namely a processor 10, atransmitter section 20, an antenna structure 30, a receiver structure 40and a display structure 50. The processor 10 generates pulse initiationsignals which are passed via a digital bus 11 to the transmitter section20. The processor also generates signals for controlling the receiversection 40, which are passed from the processor 10 to the receiversection 40 via a second digital signal bus 12.

The pulse initiation signals from the processor 10 are received at apulse duration unit 21 in the transmitter section 20, which determinesthe pulse width of the pulses to be generated. The pulses are initiatedby an edge of the pulse initiation signal, and their duration is thusfixed. The resulting pulse information is passed to a modulator 22 whichdrives a magnetron transmitter 23. That magnetron transmitter 23 isnormally a vacuum device that produces high power microwave pulses,which will form the radar signals. Those microwave pulses are passedfrom the magnetron 23 via a band pass filter 24 which controls spuriousemissions from the magnetron 23 to a circulator 25. That circulator actsas a switching unit and, at appropriate times, passes the microwavepulse to an antenna 31, from which they are transmitted. The antenna 31is arranged to rotate, and has a rotary joint 32 and a motor 33 whichdrives the antenna at a predetermined rotation speed. The motor 33 isdriven from a drive 34 which is powered from the modulator 22. Therotary joint 32 acts as a microwave connection between the rotatingantenna and the circulator 25.

When return signals are received at the antenna 31, they are passed viathe circulator 25 to a low noise converter 41 which converts the signalsto an appropriate frequency. Generally, the magnetron will producepulses in the X-band region sent on 9.4 GHz, in which case the low noiseconverter 41 will convert the received X-band signals to an IFfrequency, such as 60 MHz. Note that the circulator 25 switches betweenthe pulses for transmission from the magnetron 23 and the receivedsignals received by the antenna 31 which are passed to the low noiseconverter 41. The signals from the low noise converter 41 are passed toa PIN diode attenuator 42 which is controlled by a Time Varying Gain(TVG) generator 43 which is controlled by the processor 10 on the basisof the signals passed via bus 12. That TVG generator 43 controls thegain of the receiver section 40 to compensate for range variation of thesignal received by the antenna 31. The TVG generator 43 also controls avariable amplifier 44 which receives the output of the PIN attenuator 42and controls the IF gain of the received signal. The output of thevariable gain amplifier 44 is passed to a log detector 45 whichgenerates an output which is the logarithm of the envelope with areceived signal. That output is passed to a selectable band filter(video filter 46) which generates the output to the display section 50.

As illustrated in FIG. 1, the display section contains multiple displaystructures, each of which comprises a comparator 51, a spoke buffer 52,a signal processor 53 and a graphical display 54. However, in thearrangement shown in FIG. 1, all but one of those multiple displaystructures is mostly redundant. Thus, in FIG. 1, the output of the videofilter 46 is received by a first display structure 55, comprisingcomparator 51, spoke buffer 52, signal processor 53 and display unit 54.The comparator 51 generates a digital output that changes when the inputsignal crosses a predefined voltage threshold. The spoke buffer 52 thenreceives the output of the comparator 51, and stores a digitalrepresentation of the signal received as a function of time. The digitalsignals stored in the spoke buffer 52 are then processed by the signalprocessor 53 to generate a signal to the display unit 54. However, thosesignals to the display unit 54 are also passed directly to the displayunit 54 of a second display structure 56. In that display structure 56,the comparator 51, spoke buffer 52 and signal processor 53 areredundant. It would similarly be possible to provide more displaystructures operating a similar way.

In a sonar system, the structure is similar but the antenna is replacedby a sonar transducer which transmits and receives the sonar signals.That transducer does not rotate, unlike the antenna 31. Moreover, itwould normally be desirable for the magnetron 23 to be replaced by ahigh power RF pulse generator. In addition, since the velocity ofproportion of acoustic waves in water is substantially slower than radiowaves propagating through air, and the maximum range of targets to bedetected by a sonar system is normally less than the maximum range to bedetected in a radar system, the pulse repetition intervals and the pulsewidths used in a sonar system will be different from those used in aradar system.

SUMMARY OF THE INVENTION

The present invention seeks to improve known radar and/or sonar systemsand has a series of aspects which may be used independently, or incombination, in the radar or sonar system. Each aspect may provideadditional features to a basic antenna or sonar structure, whichincludes the principal features of any radar or sonar apparatus.

Thus, in the present invention the basic structure of a radar apparatusmay comprise:

a modulator for generating a sequence of pulse signals;

a transmitter for converting the sequence of pulse signals to radarsignals;

an antenna for emitting the radar signals and receiving return signals;

an amplifier for amplifying the return signals;

a switching device for switchedly interconnecting the transmitter to theantenna and the antenna to the amplifier; and

a signal processor processing the amplified return signals for displayand/or analysis.

Similarly, the basic structure of a sonar apparatus may comprise:

a modulator for generating a sequence of pulse signals;

a transmitter for converting the sequence of pulse signals to sonarsignals;

a transducer for emitting the sonar signals and receiving returnsignals;

an amplifier for amplifying the return signals;

a switching device for switchedly interconnecting the transmitter to thetransducer and the transducer to the amplifier; and

a signal processor for processing the amplified return signals fordisplay and/or analysis.

It can thus be seen that the basic structure of the sonar apparatusdiffers from the basic structure of the radar apparatus in thereplacement of the antenna (normally a rotating antenna) of the radarapparatus by a transducer.

The first aspect of the present invention proposes that the signalreceived by the antenna or transducer of the radar or sonar apparatusis, after amplification, subject to linear demodulation. The resultingdemodulated signal is then used to generate a display or may be analysedfurther.

Thus, this first aspect may provide a radar or sonar apparatus with thebasic structure discussed above, having a linear receiver between theamplifier and the signal processor, the linear receiver including alinear demodulator generating a digital output representing saidamplified return signals for processing by the signal processor.

A linear receiver is normally a receiver comprising linear amplifiersand linear demodulator. The output signal is linearly proportional tothe amplitude of the incoming signal. Non-linear receivers include, forexample, logarithmic and square law demodulators. Similarly, a lineardemodulator is normally a demodulator that produces an output signal, ineither digital or analog form, that is linearly proportional to theamplitude of the incoming signal. It may also provide a signal thatshows the phase of the incoming signal, with timing referenced to alocal oscillator.

The second aspect of the invention also considers the signals receivedby the antenna or transducer of the radar or sonar, and proposes thatthose signals after amplification are filtered using an anti-aliasingfilter and converted from analog to digital format via a sub-samplingconverter.

Thus, in the second aspect, there may be provided a radar or sonarapparatus with the basic structure discussed above, having ananti-aliasing filter connected to the amplifier for filtering theamplified return signals and sub-sampling analog-to-digital converterfor converting the filtered digital signals to digital signals, saiddigital signals being supplied to the signal processor.

An anti-aliasing filter is a device which is normally used before ananalog-to-digital converter, and permits the bandwidth of a signal to berestricted approximately to satisfy the Shannon-Nyquist-Kotelnikovsampling theorem. The theory requires that exact reconstruction of acontinuous-time baseband signal from its samples is possible if thesignal is band-limited and the sampling frequency is greater than twicethe signal bandwidth. In radar and sonar applications, the out of bandsignal primarily comprises white, thermal noise. To achieve asatisfactory signal to noise ratio, and provide an adequateapproximation to the above criteria, the in-band signals should not beattenuated by more than 6 dB and the potentially aliased signals beattenuated in excess of 30 dB.

As mentioned above, the first and second aspects may be independent.However, where they are used in combination, the anti-aliasing filterand the analog-to-digital converter are between the amplifier and thelinear receiver.

In the second aspect, the analog-to-digital converter produces a digitaloutput suitable for down conversion to base band or for further digitalsignal processing. For example, the digital signal may be subject todigital filtering prior to be output to the signal processor. Where thepulses thus filtered have multiple pulse widths, the filtering may be inthe form of matched filters.

Preferably, the analog-to-digital conversion, intermediate frequency(IF) filtering, and down conversion to base band, if used, are carriedout in a single integrated circuit.

Preferably, the signal is subject to non-linear dynamic range matchingprior to output. Non-linear dynamic range matching to any displayfollows the signal processing, not precedes it. The dynamic range of theradar signal, following signal processing, is still larger than thedynamic intensity range of the display. Dynamic range compression isused, but following the signal processing. A choice of algorithms forcompression is then feasible, because the compression is performed inthe signal processor. These can be log, square root, or in specialcases, linear for no compression.

The third aspect of the invention is concerned with the signalprocessor. In this third aspect, at its most general, the signalprocessor is a digital one with a network stack running on that digitalprocessor. This is to be contrasted with an arrangement in which anetwork stack runs on a separate communications processor.

Thus, the third aspect of the invention may provide a radar or sonarapparatus with the basic structure discussed above, having ananti-aliasing filter connected to the amplifier for filtering theamplified return signals, and an analog-to-digital converter forconverting the filtered signals to digital signals, said digital signalsbeing supplied to the signal processor, wherein the signal processor isa digital processor including a network stack running in the digitalprocessor.

A network stack is a software implementation of a network communicationsprotocol.

The fourth aspect of the invention also relates to the signal processor,and proposes that that processor is a digital one having a networkinterface media access controller and a physical layer interface. Thatmedia access controller may be connected directly or logically to adigital signal processor bus, possibly via bus buffers.

Thus, the fourth aspect of the invention may provide a radar or sonarapparatus with the basic structure discussed above, having ananti-aliasing filter connected to the amplifier for filtering theamplified return signals, and an analog-to-digital converter forconverting the filtered signals to digital signals, said digital signalsbeing supplied to the signal processor, wherein the signal processor isa digital signal processor having a network interface media accesscontroller and physical layer interface that are connected to a digitalsignal processing bus.

A media access controller (MAC) in a component such as an electronicintegrated circuit or functional block of a circuit that providesaddressing and access control mechanisms that makes it possible forseveral network nodes to communicate within a multipoint communicationsnetwork.

A physical layer interface (PHY) refers to an electronic integratedcircuit or functional block of a circuit interfacing between digitalsignals, and a modulation in the analog domain of a communicationsnetwork.

As mentioned above, it is known to provide a radar or sonar system inwhich different pulses are used to generate different images, e.g.images at different ranges. However, in general, the known systems haveconsidered the pulses needed for the different ranges separately, sothat one or other of the images is based on data that is significantlyolder than the data used to form the other image. The fifth aspect ofthe invention seeks to improve such arrangements, by interleaving thepulses for different ranges, which differ in pulse width and/orinter-pulse spacing, in which there is simultaneous display of images.In particular, the pulses are such that the data to form the displayimages are derived from data acquired in a duration not more than twentytimes the longer of the inter-pulse periods of the different pulses.Alternatively, at least for a radar system, the data should not beacquired over a period longer than one half the antenna revolution time,as compared with the data for the other image. The latter is notapplicable to a sonar, which does not have a rotating antenna.

Thus, in the fifth aspect of the invention, there may be provided aradar system with the basic structure discussed above, having an antennadrive for rotating the antenna with a predetermined revolution duration;

wherein the modulator is arranged to generate the sequence of pulsesignals such that the sequence comprises pulses of a first pulse patternand pulses of a second pulse pattern, the first and second pulsepatterns differing in pulse duration and inter-pulse spacing, the firstand second pulse patterns being interleaved such that within saidpredetermined revolution duration there are a plurality of pulses ofeach pulse type, and

wherein the signal processor is arranged to generate a first radar imagefrom said pulses of said first pulse type and a second radar image fromsaid pulses of said second pulse type, such that the pulses forming eachof said first and second images at any time are within a time notgreater than one half of the predetermined revolution duration.

Similarly, this fifth aspect may provide a sonar system with the basicstructure discussed above, wherein the modulator is arranged to generatethe sequence of pulses and that the sequence comprises pulses of a firstpulse pattern and pulses of a second pulse pattern, the first and secondpulse patterns differing in pulse duration and pulse spacing, the firstand second pulse patterns being interleaved and,

wherein the signal processor is arranged to generate a first sonar imagefrom said pulses of said first pulse type and a second sonar image ofsaid pulses of said second pulse type, and that the pulses forming saidfirst and second images at any time are within a time not greater thantwenty times whichever is longer of the pulse spacing of the first andsecond pulse patterns.

The sixth aspect of the invention concerns the amplifier which amplifiesthe signals received by the antenna or transducer of the radar or sonarapparatus. In the sixth aspect, a look-up table is used to control theamplifier digitally.

Thus, in the sixth aspect, there may be provided a radar or sonar systemwith the basic structure discussed above, having a controller forcontrolling the gain of the amplifier, the controller including at leastone look-up table containing data for compensating for range dependentvariation of said return signals.

Normally, the range dependent variations for radar signals is 1/R^4.However, this range dependency can depart from this function if there israin or sea-clutter, so the variation needed is often a polynomial ofrange, or a combination of range dependent power laws. Thus, if the gaincontrol of the receiver is non-linear, the look-up table contents willthen be the product of the required functions.

Where the sixth aspect is used for a radar operating at multiple ranges,whether by use of the fifth aspect of the invention or some other way,control of the amplifier will need to take into account the multipleranges. This can be done by the use of two or more look-up tables (thenumber depending on the number of ranges). Alternatively, however, thedifferent ranges may be accommodated within a single look-up table,having different table regions corresponding to the different ranges.Then, the addresses used to acquire the data for controlling theamplifier will point to the required portion of the table, depending onthe range at which the radar or sonar is currently operating. The use ofa single look-up table with different table regions will normally bepreferred.

Moreover, the use of multiple look-up tables, or a look-up table withmultiple table regions, also permits updating. Thus, when data in onetable or table region is being used, data in another table or tableregion may be up dated without affecting the current operation of theapparatus.

Where the amplifier has more than one gain stage, separate gainfunctions may be required for the stages. In this case, it is againpossible to make use of multiple look-up tables, or to having a look-uptable with different table regions for the different gain stages. Whereseparate look-up tables are used for gain control, the signal may bekept with the desired dynamic range by adding the outputs from the twotypes of tables, using saturation logic that keeps the signals used tocontrol the amplifier within the gain control range. For example, arange of 0 to 255 may be used for an eight-bit gain control.

All the aspects of the invention discussed above are applicable toeither radar or sonar systems. The seventh aspect of the invention isconcerned with a radar system, and propose that the digital signalprocessor of the basic radar structure discussed above generates adigital output which comprises greater than eight levels of radar video.Preferably, a pseudo-colour representation of the targets in amplitudeis displayed on the radar display. Preferably, there are more thansixteen levels of radar video.

With the present invention it is possible to produce a scanner withimproved display of pulse radar or sonar data. The invention may makeuse of multi-bit digital signal processing with linear conversion tobaseband. If digital control is used, this enables the scanner to combatthe effects of temperature and ageing that affect analog circuitry.

BRIEF DESCRIPTION OF THE DRAWINGS

An embodiment of the present invention will now be described in detail,by way of example, with reference to the accompanying drawings in which:

FIG. 1 is a block diagram of a known radar apparatus, and has alreadybeen described;

FIG. 2 is a block diagram of a radar apparatus embodying the presentinvention;

FIG. 3 is a graph of the relationship between gain and sensitivity of anamplifier against range;

FIG. 4 is a graph showing the frequency characteristic of a filter ofthe embodiment to FIG. 2;

FIG. 5 is a block diagram showing part of the apparatus of FIG. 2;

FIG. 6 illustrates down-conversion of signals in the embodiment of FIG.2;

FIG. 7 illustrates data buffering in the embodiment of FIG. 2;

FIG. 8 illustrates the time varying gain generator in FIG. 2;

FIG. 9 illustrates the typical characteristic of the PIN diodeattenuator shown in FIG. 2;

FIG. 10 shows pulse patterns which may be used in the embodiment of thepresent invention;

FIG. 11 is a block diagram of a known sonar system, and is similar toFIG. 1; and

FIG. 12 is a block diagram of a sonar apparatus embodying the presentinvention, and is similar to FIG. 2.

DETAILED DESCRIPTION

A radar apparatus embodying the various aspects of the invention willnow be described in detail. FIG. 2 shows the general structure of theapparatus of this embodiment. As in the known arrangement of FIG. 1, theapparatus comprises five components, namely a control processor 100, atransmitter section 200, an antenna section 300, a receiver section 400,and a display section 500. Some of the sub-components of the transmittersection 200, antenna section 300, and receiver section 400 correspond tocomponents of the transmitter section 20, the antenna section 30 and thereceiver section 40 of the arrangement of FIG. 1, and the same referencenumerals will be used for corresponding parts.

However, the apparatus of FIG. 2 is intended to generate multipledisplays at different radar ranges. Thus, the control processor 100generates two types of pulse repetition frequency signals which aretransmitted via separate digital buses 101, 102 to separate pulseduration units 201, 202. Each of those pulse duration units 201, 202determines the duration of the respected pulses, in a manner similar topulse duration unit 21. However, the duration of the pulses generated bythe pulse duration units 201, 202 will be different. The resultingsignals are combined by a logical OR component 203 before being passedto the modulator 22. As will be described in more detail later, thepulses are generated so that they are interleaved, with the manner ofinterleaving being determined by the desired ranges of images to bedisplayed.

In the arrangement of FIG. 2, the structure of the antenna section 300is similar to the antenna section 30 of the arrangement of FIG. 1. As inthe arrangement of FIG. 1, signals received by the antenna 31 are passedvia the circulator, a low noise converter (LNC) 41, a PIN attenuator 42to a variable gain amplifier 44. However, the output of that variablegain amplifier 44 is processed in a different way from the arrangementof FIG. 1, as will be described in more detail later.

In the embodiment of FIG. 2, the time varying gain (TVG) generator 401is implemented using a series of combination of fixed and variable gainamplifiers and a PIN diode attenuator. Again, the TVG generator 401 iscontrolled using signals from the control processor 100 via a digitalbus 12. However, in this embodiment, the output of the TVG generator 401is converted by respective digital-to-analog converters 402, 403 tocontrol the PIN attenuator 42 and the variable gain amplifier 44respectively.

FIG. 3 then illustrates the variation in gain sensitivity of thevariable gain amplifier 44 which is needed for different ranges. Theseare controlled by the TVG generator 401.

A fixed frequency anti-alias filter 404 follows the variable gainamplifier 44 to restrict the input signal bandwidth to:Anti-alias filter bandwidth=IF frequency+/−BW_(max).where BW_(max) is the bandwidth required for the optimum reception ofthe shortest pulses and IF is the desired intermediate frequency. Theanti-aliasing filter 404, as above, cannot have an infinite roll-offrate. However it is possible to let some signal through outside thisrange, including that which is aliased. This signal will comprisethermal (white) noise, interference and higher harmonics of the receivedtarget signal. The subsequent digital filtering will remove most of thealiased, distorted signal, when used with narrower bandwidths (longerpulses) when the lowest noise and best performance is required.Interference from other pulsed radars is rejected, even when aliasedinto the selected bandwidth. A correlation technique is used based onmultiple radar pulses. The remaining white noise, aliased into theselected bandwidth, simply degrades the noise performance marginallyaccording to the ratio of selected bandwidth to aliasing filter skirt.

The radar receiver described may use a single integrated circuit 14 bitADC with integrated high-speed digital filter. The small size of thiscomponent and the high level of integration reduce the noise coupledinto the low-noise front-end of the receiver.

FIG. 4 then illustrates the frequency response of the anti-alias filter404.

The output of the anti alias filter 404 is passed via ananalog-to-digital converter 405 to respective digital filters 406, 407for each range of the radar. Those digital filters 406, 407 receivessynchronising signals from the control processor 100 via a digital bus103. The outputs of those digital filters 406, 407 are passed viarespective spoke buffers 408, 409 to a signal processor 410, whichgenerates a network output via network unit 411 to the display section500.

FIG. 5 shows part of the apparatus of FIG. 2 in more detail. As shown inFIG. 5, the signal processor 410 comprises a digital processor (DSP)470, a static random access memory (SRAM) 471, a flash memory 472, aserial EEPROM 473 and serial digital-to-analog converter 474.

FIG. 5 also shows a field programmable gate array (FPGA) 412 in whichthe TVG generator 401 and buffers 408, 409 are embodied. In fact,although shown as a separate component in FIG. 5, the digital filter 406(and the digital filter 407) may also be embodied within the FPGA 412.

FIG. 5 also shows that the network unit 410 comprises a host portedmedia access controller (MAC) and physical layer interface (PHY) 413,and an Ethernet connector 414.

FIG. 5 also shows a sequencer 415 which controls the timing of thevarious components.

The FPGA 412 generates an output on line 475 which is passed to thecontrol processor.

This embodiment uses a sub-sampling approach to reduce theanalog-to-digital conversion sample rate. To facilitate sub-sampling,the IF frequency has been raised to 70 MHz. Without sub-sampling, theanalog-to-digital conversion would require a minimum sample rate of2*(70 MHz+(BW/2)) to satisfy the Nyquist criteria. For a 75 ns pulse,this equates to an analog-to-digital conversion sample rate of2*76.6=153.2 MHz. High resolution analog-to-digital conversions workingat this sample rate are expensive and may be subject to exportrestrictions. In practice the analog anti-alias filter 404 cannot havean infinite roll-off in the frequency domain, so the sample rate wouldneed to be somewhat higher to avoid aliasing the signal in the skirt ofthe filter.

The embodiment described uses a 57 MHz sample rate. The under-samplingin the analog-to-digital conversion aliases the signal to: 13MHz+/−BW/2. This sub-sampled signal is mixed in a complex mixer, alsopart of the digital filter IC, to produce a complex I,Q baseband signal.The complex base-band signal is then filtered in the digital filter toproduce the matched bandwidth required of 0 Hz+/−BW/2 (The mixed signalis complex I,Q). Under-sampling reduces the cost of theanalog-to-digital conversion and the processing and memory required.

When a large number of pulse-widths (say 8) is used, providing matchedanalog filtering becomes onerous to set-up, and liable to drift. Thisembodiment uses digital filtering of the IF, prior to conversion tobase-band. Such digital filtering is linear and optimal; digitising thenfiltering conventional base-band log video cannot achieve the sameresult. In this design the entire digital signal processing, takes placein the linear domain, for which many processing algorithms are available(e.g. Fast Fourier Transform (FFT))

The digital filter 406 can be re-loaded with different parameters whenthe user changes instrumented range, such that the optimised matchedbandwidth is always used on each instrumented range. Thus there is norestriction on the number of different pulse-widths that can beoptimally filtered. Alternatively, two or more filters can be madesimultaneously available and the output of these selected, on apulse-by-pulse basis, according to the required range. These filters canbe reloaded separately when the user changes one of the instrumentedranges.

FIG. 6 then illustrates the down-conversion from IF to baseband. Theoutput of the digital filter 406 is in the form of complex I,Q pairs ofsample data, which are converted into magnitude form and buffered in theFPGA 412. This buffer is illustrated in more detail in FIG. 7 in whichrespective dual ports pulse repetition interval (PRI) buffers 420, 421controlled by respective address generators 422, 423 will receiveappropriate inputs via parallel input ports 424 and output via a linkport output 425.

The TVG generator 401 will now be described in more detail. It should benoted that the TVG generator 401 effectively comprises a plurality ofgenerators for the different ranges at which the radar apparatus isoperating. FIG. 8 illustrates the part of the structure of the TVGgenerator 401 for one such range. There will be similar structureswithin the TVG generator 401 for each range at which the radar system isto operate.

The TVG generator needs to operate in real-time and thus calculation ofthe TVG function against time is identical and repeated on eachtransmitted pulse, for the same instrumented range. In this design oneinnovation is that the TVG function is implemented digitally usinglook-up tables. The look-up table contents only require re-calculationon each instrumented range change. The recalculation is the combinationof the measured non-linearity of the gain control stages, which ismeasured at the time of manufacture, and the required range or timedependant gain function, including rain or sea-clutter curves ifrequired. The final function is scaled for the sample rate of the outputof the digital filter. These tables are loaded under software controlonly when the instrumented range is changed.

Those look-up tables are illustrated at 430 and 431 in FIG. 8. Inpractice, those look-up tables may be implemented in a single table,with different table areas for the two tables 430, 431 illustrated inFIG. 8. Moreover, where there are different TVG generator structures forthe different ranges, the look-up tables of the structures for therespective ranges may themselves be combined in a single look-up table,again with different table areas for the different structures.

The loading of the lookup tables in this design is simplified by the useof dual ported SRAMs. These have a second address and data bus that isconnected to the microprocessor bus that does not interfere with readingof the lookup tables by the hardware described below. The loading of thetables and error checking if required takes place over this second dataand address bus in the same way that normal SRAM is accessed by amicroprocessor. The loading of the tables is ignored for the remainderof this discussion.

One gain control stage is normally inadequate to deal with the dynamicarrange of the input signal and preserve linearity in each of the analogstages. in this embodiment a PIN diode attenuator 42 is used before thevariable gain amplifier (VCA) 44, to prevent saturation of the input ofthe VGA 44 for large signals. This gain also needs to be varied withrange. Thus the gain control circuit should modify the gain of both thePIN diode attenuator 42 and the VGA throughout the acquisition of asingle spoke.

The PIN diode attenuator 42 is heavily non-linear with a control voltageas illustrated in FIG. 9; thus the lookup table contents for the PINdiode gain control apply the inverse non-linearity to transform thelinearly increasing (with range) lookup table address to be themultiplication of the required range dependant gain (typically powergain of R^4) and the inverse non-linear function of the PIN diodeattenuator.

In FIG. 8, the signal PRI_PLS received on line 436 synchronises thecircuit to the start of reception. A delay counter 432 delays the startof the range dependant gain control circuit to cancel the delay in thetransmitter. The delay required is loaded from the TVG delay register437 The read address counter 433 then increments, under control of clockTs, as time progresses to access the lookup table contents appropriatefor that range. In this embodiment the data for both digital-to-analogconverters 402, is interleaved, with a multiplexor, over a single 8-bitRDAC bus, the multiplexing normally occurring at a rate faster than theoutput sample rate of the digital filters.

A multiplexor 438 selects, using the digital-to-analog select signalreceived on line 439, the output of the appropriate lookup table 430 or431 dependant on which digital-to-analog converter is to be driven, thedigital-to-analog 402 connected to the PIN diode 42, or thedigital-to-analog 403 connected to the variable gain amplifier 44.

In a multi-range system, the lookup tables 430 and 431 are made largeenough to accommodate the required tables for each of the ranges, andthe most significant bits of the table read address are modified todetermine which table is to be accessed for the current range to beacquired. The range to be acquired is signalled by the control processor100 using digital bus 103.

The delay counter 432 is decremented and read address counter 433 isincremented at a rate that can be modified dependant on the output rateof the digital filter, rather than the sample rate of theanalog-to-digital converter. This sample rate is not externallyaccessible for the preferred digital filter used (AD6654); hence it isre-created by division of the analog-to-digital converter clock toproduce a clock (Ts) at this rate. Rate divider circuitry 440, delayregisters 441, rate divider phase 442 and frequency registers 443synchronise the clock Ts to commence in synchronism with signal PRI_PLSand to recur at the sample rate, add sample phase applicable to thatrange. Thus the rate divider and registers are duplicated for each rangethat may be simultaneously acquired (one of two for dual ranging systemsare shown in FIG. 8).

The use of the lower rate, Ts, reduces the number of entries required inthe range dependant table. The number of lookup table entries wouldtypically be at least as long as the maximum samples in the digitalfilter output for the range to be acquired. In practice it is possibleto reduce the number of entries. The analog output of the gaindigital-to-analog converter 434 is filtered in an analog filter, whichprovides interpolation. Alternatively, the matching of the rangedependant gain function to the desired function can be a stepwiseapproximation, provided the resulting gain steps are small and may befiltered in a later digital process before the final display.

Simultaneously to range dependant gain control, another lookup table(EXP lookup data) 435 is accessed using the EXP bits as address. Thesebits are sourced from an AGC circuit that varies the gain of the analogstages to keep the signal within the dynamic range of theanalog-to-digital converter. Three bits of EXP are available from theanalog-to-digital converter used in this embodiment. Thus in thisimplementation 8 entries are required in the EXP lookup table 435. TheEXP lookup table 435 can be used to control the gain or the EXP tablecan be set to all zeros to disable this function. Likewise if theanalogue gain control circuit is non-linear, then the EXP lookup table435 can contain the inverse non-linearity, to make the actual gain varylinearly with the value of the EXP bits. The slope of the thuslinearised function can be varied to accommodate different scaling. Forexample the analog-to-digital converter used expects the gain to vary by6 dB on each increment of EXP.

The analog-to-digital converter used in this implementation, providesEXP outputs to request a gain change in the external analog stages:

The gain in the gain-ranging block (external) is compensated for byrelinearizing, using the exponent bits EXP[2:0] of the input port. Forthis purpose, the gain control bits are connected to the EXP[2:0] bits,providing an attenuation of 6.02 dB for every increase in the gaincontrol output. After the gain in the external gain-ranging block andthe attenuation in the AD6654 (using EXP bits), the signal gain isessentially unchanged. The only change is the increase in the dynamicrange of the analog-to-digital converter.

The outputs of the selected lookup table 430 or 431 and the AGC lookuptables are added together with saturating logic that ensures the resultis kept within the range of the analog-to-digital converter and no overor underflow occurs. The contents of a diagnostic register,RDAC_DEBUG_DATA are added in this path to facilitate testing of the TVGgenerator 402.

All of the lookup table contents are twos complement numbers that can beadded together with the correct result given when negative numbersoccur. To ensure the correct timing relationship of thedigital-to-analog conversion data and control signals, the RDAC data isresynchronised in register 434 prior to leaving the TVG generator 402.

The instantaneous signal, at a particular range, can still vary due toRCS variation and fading. The overall dynamic range of received radarsignals is very large. It comprises the following elements:

Range: The received signal power for a point reflector follows the law1/R^4, (Ref. 1) where R is range. From the minimum range of say 50 m toa range of 20 NM, the dynamic range is 115 dB.

The radar cross-section (RCS) of targets of interest varies by 50 dB

Fading and multi-path effects contribute a further 30 dB of variation

Thus the total dynamic range of interest is the sum of these: 195 dB

Fortunately the range variation is predictable. TVG is applied to cancelthe effect of the range variation, subject to the limits of thermalnoise and variable gain amplifier noise and dynamic range.

The instantaneous signal, at a particular range, may vary due to radarcross section (RCS) variation and fading. In practice there is arequirement for 80 dB of instantaneous dynamic range. The theoreticaldynamic range of a 14 bit ADC is 84 dB. Signal processing, includingAzimuth integration, is used to retrieve signals below one leastsignificant bit (LSB), to further extend the dynamic range. Thus withfull range compensation using TVG, adequate instantaneous dynamic rangeexists, using such a converter, for fully linear IF processing.

The dynamic range of the radar signal, following signal processing, isstill larger than the dynamic intensity range of the display. Dynamicrange compression is used, but following the signal processing. A choiceof algorithms for compression is then feasible, because the compressionis performed in the DSP. These can be log, square root, or in specialcases, linear for no compression.

The output data comprises 8 bits (256 levels) that is converted topseudo-colour in the display's processor.

To facilitate connection of multiple displays, Ethernet is used as theconnection from the scanner to the displays.

The Ethernet media access controller/physical layer interface (MAC/PHY)connects to the bus of the digital signal processor and the networksoftware stack runs on the digital signal processor itself, not aseparate communications processor. This saves the additional cost andcomplexity of a multi-processor architecture.

FIG. 2 also shows that the display section 500 of the radar apparatusgenerates different displays in multiple windows 501, 502, 503 (thetotal number n of such windows depending on the number of ranges at thewhich the radar is operated). Each window 501, 502, 503 receives adigital signal from the network unit 411 and displays an imagescorresponding to one of the ranges of the radar.

Known multiple range radars interleave target detection of ranges on arevolution-by-revolution basis. The radar of this embodiment interleavesdetection of targets on different instrumented ranges, and the displayof the data for these ranges on a pulse-by-pulse basis. Thus there issimultaneous display of radar images from multiple instrumented rangesthat was acquired at virtually the same instant, and is fullytime-locked. The images are updated in real time so that the images arenot displaying radar data older than that from a fraction of arevolution of the antenna.

Interleaving of radar pulses of different characteristics is known, butthe information from the different pulses is used to create separate,simultaneous displays for different instrumented ranges.

Radar pulse characteristics including pulse width and repetition ratehave to be optimised for range discrimination or signal to noise ratiodependent on the range to be displayed. Likewise pulses cannot betransmitted until the echoes for that range have been received. Theprocessing of the received signals is filtered to optimise reception foreach of the pulses characteristics. Pulses of different types areinterleaved on transmission and the targets' echoes received are passedalong separate paths. In the receiver each path is separately optimisedfor that pulse shape. More than one pulse of each type can betransmitted as part of the pulse pattern to permit the higher repetitionrates required at shorter ranges. The pulse pattern is arranged to givethe required ratio of pulse repetition rates for each range such thatthe transmitter's duty cycle is not exceeded.

A suitable pulse pattern is illustrated in FIG. 10. In FIG. 10 pulses ofpulse width P1 is the pulse duration for pulses optimised for a firstrange, and T1 is the acquisition interval for targets at that range.Similarly, P2 is the pulse duration for pulses optimised for a secondrange, with T2 being the corresponding acquisition interval. In eachcase, the acquisition interval follows the corresponding pulse, but isseparated from the immediately following pulse by a variableinterference rejection inter-pulse jitter period Δ. This de-correlatestargets at each range. In such an arrangement, the pulse repetitioninterval is the sum of the pulse duration, the acquisition interval andthe maximum value of Δ for each range.

FIG. 10 shows that the pulses are interleaved, and in this embodimentthere are three pulses for the second range between each pulse for thefirst range. Other combinations are, however, possible. However, table 1then illustrates a possible pulse pattern, where (a, b) is the number ofpulses of each type that are interleaved from (column a, row b).

TABLE 1 Pulse width (seconds) Pattern 75.0E−9 100.0E−9 150.0E−9 250.0E−9350.0E−9 450.0E−9 600.0E−9 1.0E−6 Range (1) Range (2) Nominal PRI (sec)Pulse width Nominal PRI 333.3E−6 333.3E−6 333.3E−6 333.3E−6 500.0E−6625.0E−6 769.2E−6 1.2E−3 (seconds) (sec) Mode 1 2 3 4 5 6 7 8  75.0E−9333.3E−6 1 1,1 1,1 1,1 1,1 3,2 2,1 7,3 11,3 100.0E−9 333.3E−6 2 0,0 1,11,1 1,1 3,2 2,1 7,3 11,3 150.0E−9 333.3E−6 3 0,0 0,0 1,1 1,1 3,2 2,1 7,311,3 250.0E−9 333.3E−6 4 0,0 0,0 0,0 1,1 3,2 2,1 7,3 11,3 350.0E−9500.0E−6 5 0,0 0,0 0,0 0,0 1,1 4,3 3,2  5,2 450.0E−9 625.0E−6 6 0,0 0,00,0 0,0 0,0 1,1 7,6 11,6 600.0E−9 769.2E−6 7 0,0 0,0 0,0 0,0 0,0 0,0 1,1 8,5  1.0E−6  1.2E−3 8 0,0 0,0 0,0 0,0 0,0 0,0 0,0  1,1These Patterns are Calculated from the Ratio of PRIs for Each Row orColumn.

In general, the pulse repetition rate (PRI) needs to be sufficient thatenough pulses are generated within a suitable time to generate theappropriate displays simultaneously. In general, the pulse repetitionmust be sufficient that the data in any image is acquired over a timewhich is not greater than twenty times the longest interpulse period, inthe case of FIG. 10 being the period between each pulse P1 for a radar,this duration also needs to be longer than one half of the antennaresolution time. Thus, in the embodiment of FIG. 2, it is necessary thatthe pulse repetition frequency determined by the control processor 100is related to the speed of the motor 33. This may be pre-set, or couldbe based on measurement.

When used with radar, the magnetron transmitter 23 can be replaced byanother source of microwaves, such as an oscillator, and high powertravelling wave tube amplifier or lower power solid state amplifier. Inboth sonar and radar, if lower power transmitters are used, it is normalto employ pulse compression to increase the average power transmitted,whilst not reducing the range resolution. The linear receiver used inthis is suitable for use with radar and sonar employing pulsecompression.

The above embodiment illustrates a radar apparatus. A sonar systemembodying the present invention may be similar, although with theantenna 31 replaced by a transducer, and with the magnetron replaced bya high power RF generator. Moreover, the pulse intervals and pulsewidths of Table 1 would then be modified to accommodate the slowervelocity of acoustic waves in water to that of radio waves in air.

FIG. 11 of the accompanying drawings shows a known sonar apparatus. Mostof the sonar apparatus of FIG. 11 is similar to the radar apparatus ofFIG. 1, and corresponding parts will be indicated by the same referencenumerals. Thus, the sonar apparatus of FIG. 11 comprises five principalcomponents, namely a processor 10, a transmitter section 20, an antennastructure 603, a receiver structure 40 and a display structure 50. Theprocessor 10 generates pulse initiation signals which are passed via adigital bus 11 to the transmitter section 20. The processor alsogenerates signals for controlling the receiver section 40, which arepassed from the processor 10 to the receiver section 40 via a seconddigital signal bus 12. The display structure 50 of this sonar apparatusis the same as the display structure 50 of the radar apparatus of FIG. 1and thus will not be described in detail now.

The pulse Initiation signals from the processor 10 are received at apulse duration unit 21 in the transmitter section 20, which determinesthe pulse width of the pulses to be generated. The pulses are initiatedby an edge of the pulse initiation signal, and their duration is thusfixed. This is the same as in the radar apparatus of FIG. 1 however, theresulting pulse information is passed to a high power amplifier 601which drives a transformer 602 to match the oscillator to the sonartransducer 603. That transformer 602 passes the sonar pulse to thetransducer 603 from which they are transmitted. That sonar transducer603 is normally a piezo-electric device that produces high poweracoustic pulses in water, which will form the sonar signals. When returnsignals are received at the transducer 603, they are passed via thetransformer 602 to a low noise amplifier 604 which increases theamplitude of the signals to an appropriate level. Band-pass filter 605filters the received signals to reduce unwanted noise. Generally, thehigh power oscillator will produce pulses in the Long and Mediumwave-band region 50 to 250 KHz, in which case the band-pass filter willbe tuned to the appropriate centre frequency.

The signals from the band pass filter 605 are passed to an attenuator606 which, in a similar way to the PIN alternator 42 of FIG. 1, iscontrolled by a Time Varying Gain (TVG) generator 43 which is Controlledby the processor 10 on the basis of the signals passed via a bus 12.That TVG generator 43 controls the gain of the receiver section 40 tocompensate for range variation of the signal received by the transducer.The TVG generator 43 also controls a variable amplifier 44 whichreceives the output of the attenuator 606 and controls the gain of thereceived signal. The output of the variable gain amplifier 44 is passedto a log detector 45 which generates an output which is the logarithm ofthe envelope with a received signal. That output is passed to aselectable band filter (video filter 46) which generates the output tothe display section 50.

A sonar apparatus embodying the various aspects of the invention willnow be described in detail. FIG. 12 shows the general structure of theapparatus of this embodiment. Again, most of the sonar apparatus of FIG.12 is similar to the radar apparatus of FIG. 2, and corresponding partsare indicated by the same reference numerals. The sonar apparatus ofFIG. 12 also has many of the features of the sonar apparatus of FIG. 11,and again corresponding parts are indicated by the same referencenumerals. Thus, the apparatus comprises five components, namely acontrol processor 100, a transmitter section 200, a transducer section603, a receiver section 400, and a display section 500. The displaysection of the sonar apparatus of FIG. 11 is the same as the displaysection 500 of the radar apparatus of FIG. 2, and so will not bedescribed in detail now.

Unlike the sonar apparatus of FIG. 11, the apparatus of FIG. 12 isintended to generate multiple displays at different sonar ranges. Thus,the control processor 100 generates two types of pulse repetitionfrequency signals which are transmitted via separate digital buses 101,102 to separate pulse duration units 201, 202. Each of those pulseduration units 201, 202 determines the duration of the respected pulses,in a manner similar to pulse duration unit 21. However, the duration ofthe pulses generated by the pulse duration units 201, 202 will bedifferent. The resulting signals are combined by a logical OR component203 before being passed to the high power oscillator 601. The pulses aregenerated so that they are interleaved, with the manner of interleavingbeing determined by the desired ranges of images to be displayed.

In the arrangement of FIG. 12, the structure of the transducer section603 is similar to the transducer section 603 of the arrangement of FIG.11. As in the arrangement of FIG. 11, the transducer 603 receivessignals from the high power amplifier 601 via the transformer 602. Thereturn signals received by the transducer 602 are passed via thetransformer 602, a low amplifier (LNA) 604 a band-pass filter 605, anattenuator 606 to a variable gain amplifier 44. However, the output ofthat variable gain amplifier 44 is processed in a different way from thearrangement of FIG. 11, namely in the same way as the output of thevariable gain amplifier 44 is processed in the embodiment of FIG. 2.

In the embodiment of FIG. 12, the time varying gain (TVG) generator 401is implemented using a series of combination of fixed and variable gainamplifiers and an attenuator. Again, the TVG generator 401 is controlledusing signals from the control processor 100 via a digital bus 12.However, in this embodiment, the output of the TVG generator 401 isconverted by respective digital-to-analog converters 402, 403 to controlthe attenuator 606 and the variable gain amplifier 44 respectively.

As mentioned above, the output of the variable gain amplifier 44 isprocessed in the same way as the output of the variable gain amplifier44 in the embodiment of FIG. 2. The components of the receiver section400 and display section 500 which process signals from the variable gainamplifier 44 are the same in FIGS. 2 and 11. Thus, the processing of thesignals is as described with reference to FIGS. 3 to 10. Normally, therewill then be differences in the display generator, since radarapparatuses tend to generate circular display, which is not appropriatefor sonar arrangements. Also, as previously mentioned, the pulseintervals and pulse widths of Table 1 would have to be modified toaccommodate the slow velocity of acoustic waves in water, as comparedwith radio waves in air. However, these differences do not affect thesignal processing described with reference to the first embodiment.

The invention claimed is:
 1. An apparatus comprising: a modulator forgenerating a sequence of pulse signals; a transmitter for converting thesequence of pulse signals to ranging signals; an antenna for emittingthe ranging signals and receiving return signals; an amplifier foramplifying the return signals; a switching device for switchedlyinterconnecting the transmitter to the antenna and the antenna to theamplifier; and a signal processor processing the amplified returnsignals for display and/or analysis, wherein the apparatus compriseseither a radar apparatus that provides the ranging signals as radarsignals or a sonar apparatus that provides the ranging signals as sonarsignals, characterised in that: the apparatus further includes a linearreceiver between the amplifier and the signal processor, the linearreceiver including a linear demodulator generating a digital outputrepresenting the amplified return signals for processing by the signalprocessor.
 2. The apparatus of claim 1, wherein the linear receivercomprises a plurality of linear amplifiers.
 3. The apparatus of claim 1,further comprising: an analog-to-digital converter for converting theamplified return signals to digital signals; and an anti-aliasing filterconnected to the amplifier for filtering the amplified return signals,wherein the analog-to-digital converter is a sub-samplinganalog-to-digital converter for converting the filtered signals to thedigital signals.
 4. The apparatus of claim 3, wherein the anti-aliasingfilter is arranged to restrict the bandwidth of the amplified returnsignals to satisfy the Shanon-Nyquist- Kotelnikov sampling theorem. 5.The apparatus of claim 3, further comprising a digital filter forfiltering the digital signals from the sub-sampling analog to digitalconverter.
 6. The apparatus of claim 1, further comprising a displayarranged to show a pseudo-colour representation of the amplitude of thereturn signals from the output of the signal processor.
 7. The apparatusof claim 1, further comprising: an analog-to-digital converter forconverting the amplified return signals to digital signals; and ananti-aliasing filter connected to the amplifier for filtering theamplified return signals, wherein the analog-to-digital converter is ananalog-to-digital converter for converting the filtered signals to thedigital signals, wherein the signal processor is a digital processorcomprising a network stack running in the digital processor.
 8. Theapparatus of claim 1, further comprising: an analog-to-digital converterfor converting the amplified return signals to digital signals; ananti-aliasing filter connected to the amplifier for filtering theamplified return signals, wherein the analog-to-digital converter is ananalog-to-digital converter for converting the filtered signals to thedigital signals, wherein the signal processor is a digital signalprocessor; and a network interface media access controller and aphysical layer interface that are connected to a digital signalprocessing bus.
 9. The apparatus of claim 1, wherein the apparatus is aradar apparatus and the radar apparatus further comprises an antennadrive for rotating the antenna with a predetermined revolution duration;wherein the modulator is arranged to generate the sequence of pulsesignals such that the sequence comprises pulses of a first pulse patternand pulses of a second pulse pattern, the first and second pulsepatterns differing in pulse duration and inter-pulse spacing, the firstand second pulse patterns being interleaved such that within thepredetermined revolution duration there are a plurality of pulses ofeach pulse type; and wherein the signal processor is arranged togenerate first radar image from the amplified return signals associatedwith the pulses of the first pulse type and a second radar image fromthe amplified return signals associated with the pulses of the secondpulse type, such that the pulses of the first and second pulse types atany time are within a time not greater than one half of thepredetermined revolution duration.
 10. The apparatus of claim 1, whereinthe apparatus is a radar apparatus and the radar apparatus furthercomprises an antenna drive for rotating the antenna with a predeterminedrevolution duration; wherein the modulator is arranged to generate thesequence of pulse signals such that the sequence comprises pulses of afirst pulse pattern and pulses of a second pulse pattern, the first andsecond pulse patterns differing in pulse duration and inter-pulsespacing, the first and second pulse patterns being interleaved such thatwithin the predetermined revolution duration there are a plurality ofpulses of each pulse type; and wherein the signal processor is arrangedto generate a first radar image from the amplified return signalsassociated with the pulses of the first pulse type and a second radarimage from the amplified return signals associated with the pulses ofthe second pulse type, such that the pulses of the first and secondpulse types at any time are within a time not greater than twenty timeswhichever is larger of the pulse spacing of the first and second pulsepatterns.
 11. The apparatus of claim 1, wherein the apparatus is a sonarapparatus and the modulator is arranged to generate the sequence ofpulses and that the sequence comprises pulses of a first pulse patternand pulses of a second pulse pattern, the first and second pulsepatterns differing in pulse duration, pulse spacing, and beinginterleaved; and wherein the signal processor is arranged to generate afirst sonar image from the amplified return signals associated with thepulses of the first pulse type and a second sonar image of from theamplified return signals associated with the pulses of the second pulsetype, and that the pulses of the first and second pulse types at anytime are within a time not greater than twenty times whichever is longerof the pulse spacing of the first and second pulse patterns.
 12. Theapparatus of claim 1, further comprising a controller for controllingthe gain of the amplifier, the controller comprising at least onelook-up table containing data for compensating for any range-dependentvariation of the return signals.
 13. The apparatus of claim 12, whereinthe apparatus is configured to operate at multiple ranges and whereinthere are multiple look-up tables, each range having a correspondinglook-up table.
 14. The apparatus of claim 12, wherein the apparatus isconfigured to operate at multiple ranges and wherein the at least onelook-up table has multiple table regions, each range having acorresponding table region.
 15. The apparatus of claim 12, wherein theamplifier has a plurality of gain stages, and wherein there are aplurality of the look-up tables, each table corresponding to one of thegain stages.
 16. The apparatus of claim 12, wherein the amplifier has aplurality of gain stages, and wherein the at least one look-up table hasa plurality of table regions, each table region corresponding to one ofthe gain stages.
 17. The apparatus of claim 15, further comprising anadder configured to add together outputs from the plurality of look-uptables for the plurality of table regions and saturation logic arrangedto be applied to the added-together outputs, wherein an appropriate gainstage of the amplifier is determined on the basis of the saturationlogic.
 18. The apparatus of claim 1, wherein the apparatus is a radarapparatus and the signal processor is a digital signal processorarranged to generate a digital output which comprises greater than eightlevels of radar video.
 19. The apparatus of claim 18, further comprisingmeans for subjecting the digital output of the digital signal processorto non-linear dynamic range matching.
 20. The apparatus of claim 3,wherein the apparatus is a radar apparatus; wherein the ranging signalsand the received return signals are X-band signals; the apparatusfurther comprises a converter for converting the X-band return signalsto intermediate frequency return signals; wherein the amplifier operateson the intermediate frequency return signals; wherein the filteroperates on the amplified intermediate frequency return signals; andwherein the sub-sampling analog-to-digital converter samples thefiltered amplified intermediate frequency return signals at a samplerate less than the intermediate frequency.